Multi-phase converter

ABSTRACT

Disclosed is a multi-phase converter comprising a plurality of electric phases, each of which can be triggered by a switching means. At least one coupling means ( 100  to  106 ) is provided for coupling a phase to another phase, each phase having two windings.

BACKGROUND OF THE INVENTION

The invention relates to a multiphase converter.

A power converter, which can also be referred to as a DC-to-DCconverter, is designed to convert a DC voltage at the input into a DCvoltage with a different voltage level. Such a power converter can alsobe in the form of a multiphase converter with coupled inductances.Coupled multiphase converters comprise a plurality of phases, whereineach phase is routed through a conductor through which current flows,which conductors are coupled to one another by magnetic coupling means.Current ripple generated by each phase as a result of the magneticcoupling can, however, impair the operation of the multiphase converter,for which reason sections of the individual conductors need to bearranged in a suitable manner with respect to one another in order toavoid, for example, disadvantages associated with electromagneticcompatibility.

Document WO 2012/028558 A1 describes a multiphase converter in which,owing to the coupling of one phase of at least six phases to at leastthree other phases in magnetic opposition, disruptive mutual influencingof the phases is minimized and a large proportion of the magnetic fluxis compensated for. The phases to be coupled are in this case selectedsuch that optimum compensation can be achieved. This takes place by anopposing current profile of the phases. The aim here is for the phasesto be magnetically coupled in such a way that the resultant magneticfield owing to the coupled phases is minimized. In this case, a ferritecore is used for coupling the magnetic fluxes in order to profit fromthe high permeability of the material. Given the coupling proposed here,the phases can be driven in order and independently of one another.

Document DE 10 2010 040 202 A1 discloses a multiphase convertercomprising a plurality of phases, which are each drivable by a switchingmeans. In this case, the coupling means is intended to magneticallycouple at least one of the phases to at least three other phases.

Document DE 10 2010 040 222 A1 describes a multiphase converter for aplurality of electrical phases, which are each drivable by switchingmeans. In this case, coupling means are provided which magneticallycouple at least a first phase to at least a second phase, wherein atleast two phases run spatially in one plane.

A similar multiphase converter is described in document DE 10 2010 040205 A1. In said document, at least two coupling means are provided,wherein at least one of the two coupling means has a lower inductancethan the other coupling means.

Various concepts with coupled inductances are therefore known, but atleast some of these concepts have different disadvantages, of which somewill be mentioned below by way of example. For example, excessivecoupling results in EMC problems. An excessively complex design resultsin high manufacturing costs. If the phase start and the phase end arespatially separate, this results in disadvantages in respect of EMC andefficiency as a result of conductor loops. Furthermore, high core lossesduring no-load operation cause poor efficiency on a reduced load. Adesign which is excessively flat requires an excessively large amount ofinstallation space. In addition, a high volume requirement forsoft-magnetic material results in high costs.

Care should be taken to ensure that, in the power electronics, there areDC-to-DC converter topologies in which a leakage flux is requiredexplicitly for operation. These include, inter alia, converters withcoupled inductances. In special leakage transformers, such as, forexample, doorbell transformers, welding transformers, seriestransformers etc., the leakage flux is used for short-circuit withstandcapability and therefore for current regulation. This additional leakageflux can to a small extent be influenced by the type and/or embodimentof the copper winding in the ferrite core.

In the cases where this low, additionally achievable leakage inductanceis insufficient, additional inductances, for example with componentparts, need to be built into the current path. In addition to additionalinstallation space, this also results in additional costs for componentparts and power losses.

A further multiphase converter is known from WO 2009/114873 A1, forexample. The DC-to-DC converter described therein comprises a coilcomprising a nonlinear inductive resistor, a switching system and anoutput filter. In this case, adjacent phases are coupled to one another.

EP 1 145 416 B1 has already disclosed a converter for convertingelectrical energy. It is thus proposed here that the inductor size canbe reduced by using coupled inductances. In this case, the coupledinductors are intended to be dimensioned such that the load currents inthe subbranches compensate for one another and do not result in anymagnetic loading of the inductors. Only the differential current betweenthe individual subbranches then results in a magnetic field.

US 2009/0179723 A1 already discloses a DC-to-DC converter, in which theleakage inductance can be set by means of a distance between two phasesto be magnetically coupled to one another which is selected in atargeted manner.

SUMMARY OF THE INVENTION

Against this background, a multiphase converter according to theinvention and a coupling concept are proposed.

A coupling concept which is optimized in terms of installation space,costs and efficiency in respect of the requirements for automotivehigh-power converters is therefore proposed.

It is important that phases are magnetically coupled to coupling means,wherein each coupling means couples in each case at least two phases,wherein each phase comprises in each case at least two turns.

The multiphase converter can be configured in such a way that, byintroducing a magnetic leakage flux in a targeted manner, the couplingof the phases is slightly reduced, as a result of which a reduction inthe power loss and therefore an increase in the efficiency can beachieved. Thus, interference could also be further reduced. This isachieved by a means for influencing a magnetic leakage flux, which meansis arranged between at least two of the phases to be coupled. Thissolution manages without any additional installation space and istherefore space-saving.

It should be noted that the proposed multiphase converter can bedesigned for a number of phases. For example, two, three, four, five,six or more phases can be provided. At least two turns are in this caseassociated with each phase.

Particularly expediently, the means for influencing a magnetic leakageflux is connected to the coupling means in such a way as to conductmagnetic flux. Thus, this functionality could be integrated even duringmanufacture of the coupling means, for example by pressing the ferritecores as coupling means directly in the production process.

An expedient development provides that the means for influencing amagnetic leakage flux is arranged centrally between the two phases to becoupled. Particularly expediently, the means for influencing the leakageflux is rectangular or dome-shaped. An expedient development providesthat the means for influencing the leakage flux is connected to thecoupling means on only one side. This variant can be manufactured in aparticularly simple manner since the means for influencing the leakageflux is part of the coupling means and consists of the same material.

An expedient development provides that a gap, preferably an air gap,preferably of the order of magnitude of 1 mm, particularly preferablybetween 0.3 and 0.5 mm, is provided between the means for influencingthe leakage flux and the coupling means. This size avoids any undesiredsaturation effects beyond a certain tolerance band.

An expedient development provides that the at least one phase isembodied as a round wire. This increases the leakage fluxes and it isthus possible for the interference to be further reduced.

An expedient development provides that a first phase has substantially aplanar, U-shaped profile, while a second phase has a substantiallyrectangular, planar profile. These phases with such a design can besurrounded by coupling means, preferably conventional ferrite cores. Asa result, the desired coupling of the phases is achieved in a verysimple manner by using a matrix-shaped design.

A number of phases, for example two to five, can be provided. In thiscase, the coupling means couple in each case one phase to precisely oneother phase, to two phases, to three or more phases. It is expedient ifthe individual phases can still be controlled independently of oneanother. In this case, at least two turns are associated with eachphase.

A particularly expedient development provides that precisely six phasesare provided, wherein the coupling means magnetically couple each of thesix phases to three further phases of the six phases. This type ofcoupling ensures firstly that the individual phases can still becontrolled independently of one another. In addition, the failsafety ofthe multiphase converter can be increased owing to the increased levelof interconnection of the phases.

An expedient development provides that the coupling means comprises atleast two parts, wherein one of the parts has a U-shaped, O-shaped,I-shaped or E-shaped cross section. By virtue of this design, the phasesto be coupled can be surrounded by the coupling means in a particularlysimple manner. An expedient development provides that a gap, preferablyan air gap, is provided between two parts. In this way, the inductancecan be influenced particularly easily. An expedient development providesthat a plurality of coupling means comprising at least two parts have atleast one common part, preferably a metal plate. Thus, fitting could befacilitated since all of the coupling means could be closed bypositioning of the plate in only one step.

An expedient development provides that at least two, in particularthree, coupling means are provided in order to magnetically couple oneof the phases to two further phases, wherein at least one of the twocoupling means has a lower inductance than the other coupling means. Bysuitable selection of the inductance of the coupling means, variousaspects can be influenced and optimized. Firstly, the inductanceinfluences the power loss and therefore also the development of heat inthe coupling means. A reduction in the inductance also reduces the powerlosses. In addition, a lower inductance can act as protection againstsaturation. As a result, coupling means with a lower inductance onlyenter saturation later at higher currents, with the result that themultiphase converter can be operated for longer in a stable operatingstate in the event of a fault. Secondly, a high inductance reduces thecurrent ripple. Thus, by selecting the suitable inductance, thedistribution of losses, saturation response and current ripple can beoptimized.

An expedient development provides that the coupling means which couplesone phase to a phase which is driven with a phase shift substantiallythrough approximately 180° has a lower inductance than at least one ofthe other coupling means. As a result, said coupling means which aregenerally subjected to a greater load can be reduced in terms of lossessuch that a lower development of heat is also achieved.

BRIEF DESCRIPTION OF THE DRAWINGS

Further advantages and configurations of the invention result from thedescription of the attached drawings.

It goes without saying that the features mentioned above and yet to beexplained below can be used not only in the respectively specifiedcombination, but also in other combinations or on their own, withoutdeparting from the scope of the present invention.

FIG. 1 shows coupling elements which couple phases to one another.

FIG. 2 shows a coupling element in a side view.

FIG. 3 shows the coupling concept.

FIG. 4 shows further coupling elements.

FIG. 5 shows a circuit arrangement.

FIG. 6 shows a schematic illustration of the respective coupling of thephases.

FIG. 7 shows driving and current characteristics in the case of theexemplary embodiment shown in FIG. 5.

FIG. 8 shows a section through a coupling means comprising two coupledphases.

FIG. 9 shows a section along the line A indicated in FIG. 4 in the caseof a matrix-shaped design comprising nine coupling means in a plan view.

FIG. 10 shows a section through a coupling means comprising means forinfluencing the leakage flux comprising two coupled phases in analternative exemplary embodiment.

FIG. 11 shows an alternative exemplary embodiment of a coupling meanscomprising an air gap and means for influencing the leakage flux.

FIG. 12 shows a section through a coupling means comprising dome-shapedmeans for influencing the leakage flux comprising two coupled phases ina further alternative exemplary embodiment.

FIG. 13 shows a section through a coupling means comprising rectangularmeans for influencing the leakage flux comprising two coupled phases ina further alternative exemplary embodiment.

FIG. 14 shows a section through a coupling means comprising rectangularmeans for influencing the leakage flux comprising two coupled phases ina further alternative exemplary embodiment, in which first and secondphase are always arranged alternately.

FIG. 15 shows a section through a coupling means comprising means forinfluencing the leakage flux comprising two coupled phases in the formof round wires in a further alternative exemplary embodiment.

DETAILED DESCRIPTION

The invention is illustrated schematically in the drawings on the basisof embodiments and will be described in detail below with reference tothe drawings.

The figures show the compact and flat design of the four-phase system ofcoupled inductances with adjustable leakage flux.

In this system, in each case two phases are coupled to one another percoupling element or core. In the four cores, each phase is thereforemagnetically coupled to its predecessor and successor (in time). Byvirtue of the embodiment with in each case two turns per phase, themagnetic saturation of the core cross section is markedly reduced. Thisin turn results in markedly lower core losses and therefore in lesscomplexity in terms of cooling.

Owing to the fact that no air gap is required in the useful flux path incoupled systems, high inductance values can also be realized with thetwo turns. Owing to the increased inductance, any core can be providedwith an air gap as protection against saturation.

In comparison with embodiments with only one turn, the current which canbe tolerated as imbalance/load splitting between the phases is doubled.The additional segments are used to introduce the desired leakage flux.This can take place during manufacture by grinding the air gap. Thedesign is simple, and therefore stamped or bent parts can be insertedinto the lower core half and can be closed from above with an identicalsecond core half. Additional winding techniques are dispensed with.

If the installation space available does not permit a flat extent, the2×2 matrix can also be folded to form a narrow and more compact design.

This results in a plurality of advantages for low-voltage automotiveconverters having high powers and efficiencies:

1) high efficiency owing to low copper resistance,2) low quiescent losses and thus high efficiency on reduced load,3) high dynamics owing to coupling,4) easily adaptable leakage flux and thus optimizable EMC response tothe load demands,5) small conductor loop since each phase can be passed directly out ofthe core where it is merely guided. As a result, it can be terminateddirectly at the switching cell with a capacitor, which is advantageousfor EMC.

FIG. 1 shows coupling means 100, 102, 104 and 106, which each couple twophases to one another, wherein each phase comprises two turns.

FIG. 2 shows the coupling means 102 comprising a first turn 110, asecond turn 112, a third turn 114 and a fourth turn 116. The first turn110 and the second turn 112 form a first phase, and the third turn 114and the fourth turn 116 form a second phase. These two phases aremagnetically coupled to one another in the coupling means 104.

This coupling is illustrated once again in FIG. 3. It is important thatthe turns of the phases can be arranged next to one another and oneabove the other. Furthermore shown are turns 120 and 122, which form aphase, and turns 130 and 132, which form a phase.

FIG. 4 shows a further illustration of coupling means 150, 152, 154 and156, which each couple two phases, which each comprise two turns.

The design of a multiphase converter 10 is illustrated in terms ofcircuitry in FIG. 5. The multiphase converter 10 described by way ofexample here comprises six phases 11 to 16. Each of the phases 11 to 16can be driven individually via corresponding switching means 21 to 26,in each case comprising a high-side switch and a low-side switch. Eachcurrent of the phases 11 to 16 flows, owing to magnetic coupling tothree further phases, through three inductances Lxx, which magneticcoupling is effected by the corresponding coupling means 31 to 39. Afirst coupling means 31 couples the first phase 11 to the second phase12 magnetically, with the result that an inductance L12 results for thefirst phase 11, and an inductance L21 results for the second phase 12. Asixth coupling means 36 couples the first phase 11 to the sixth phase 16magnetically, with the result that an inductance L16 results for thefirst phase 11, and an inductance L61 results for the sixth phase 16. Aseventh coupling means 37 couples the first phase 11 to the fourth phase14 magnetically, with the result that an inductance L14 results for thefirst phase 11, and an inductance L41 results for the sixth phase 16. Asecond coupling means 32 couples the second phase 12 to the third phase13 magnetically, with the result that an inductance L23 results for thesecond phase 12, and an inductance L32 results for the third phase 13. Aninth coupling means 39 couples the second phase 12 to the fifth phase15 magnetically, with the result that an inductance L25 results for thesecond phase 12, and an inductance L52 results for the fifth phase 15. Athird coupling means 33 couples the third phase 13 to the fourth phase14 magnetically, with the result that an inductance L34 results for thethird phase 13, and an inductance L43 results for the fourth phase 14.An eighth coupling means 38 couples the third phase 13 to the sixthphase 16 magnetically, with the result that an inductance L36 resultsfor the third phase 13, and an inductance L63 results for the sixthphase 16. A fourth coupling means 34 couples the fourth phase 14 to thefifth phase 15 magnetically, with the result that an inductance L45results for the fourth phase 14, and an inductance L54 results for thefifth phase 15. A fifth coupling means 35 couples the fifth phase 15 tothe sixth phase 16 magnetically, with the result that an inductance L56results for the fifth phase 15, and an inductance L65 results for thesixth phase 16.

An input current IE is distributed among the six phases 11 to 16. Acapacitor as filter means is connected to ground at the input. Theoutputs of the phases 11 to 16 are combined at a common summation pointand connected to ground by means of a capacitor (not designated), asfilter means. Then, the output current IA is present at the commonoutput-side summation point. The inductances Lxx respectively coupled toone another are oriented with a different winding sense with respect toone another as indicated by the corresponding points in FIG. 1.

In this case, at least two turns or windings can be provided for eachphase.

FIG. 6 illustrates systematically how the six phases 11 to 16 arecoupled to one another by corresponding coupling means 31 to 39. Asalready described in connection with FIG. 1, both adjacent phases arecoupled to one another and also, in addition, the phase offset through180°. An adjacent phase is understood to mean one which is drivendirectly previously or subsequently in time, i.e. whose switch-on timesare directly preceding or succeeding in time. In the exemplaryembodiment, the designation of the phases 11 to 16 is selected such thatthe phases 11 to 16 are driven successively corresponding to thenumbering, i.e. in the following order (figures correspond to thereference signs of the phases): 11-12-13-14-15-16-11 etc., in each casephase-shifted through 60 degrees or through T/6 (360 degrees/number ofphases), where T is the period of a drive cycle. This order is alsoshown in FIG. 2 and FIG. 7. That is to say that the start times for thevarious phases 11 to 16 are phase-shifted through in each case 60degrees or shifted in time through in each case T/6. Admittedly therespective phase is switched off again after the time duration T/6 inFIG. 7 (PWM ratio 1/6). Depending on the desired voltage ratio, thedisconnection could be earlier or later up to permanently on Te,depending on the desired PWM signal (between 0% (permanently off, Te=0)and 100% (permanently on, Te=T), based on a period T).

The graph shown in FIG. 7 shows the time characteristics of the drivesignals 52 for the respective switching means 21 to 26 of thecorresponding phases 11 to 16 and the current characteristics in phases11 to 16. The switching means 21 to 26 energize the associated phases 11to 16 successively for in each case one sixth of a period T, for exampleby a PWM signal, and are then in the freewheeling mode. The resultantcurrent characteristics of the individual phases 11 to 16 are shown byway of example below this. The period T of the drive signals 52 is ofthe order of magnitude of 0.01 ms, for example. The start times for thedifferent phases 11 to 16 are phase-shifted through in each case 60degrees or shifted in time through T/6. The start time of the secondphase 12 with the corresponding drive signal 52 of the second switchingmeans 22 is t=0 and is switched off again after 1/6 T (depending on thedesired PWM ratio). The start time of the third phase 13 adjacent to thesecond phase 12 is T/6, the start time of the fourth phase 14 is 2T/6,and so on. Admittedly the respective phase is switched off again afterT/6 (PWM ratio 1/6) in FIG. 7. Depending on the desired voltage ratio,however, the disconnection could take place earlier or later, up topermanently on, depending on the desired PWM signal (between 0%(permanently off) and 100% (permanently on)). That is to say that aplurality of phases 11 to 16 could also be energized simultaneously at aspecific time if this is required by the desired voltage ratios. Thestart times are temporally offset, however.

FIG. 8 shows the design of the coupling means 31-39 by way of examplewith reference to the first coupling means 31, which magneticallycouples the first phase 11 and the second phase 12. The first couplingmeans 13 comprises a substantially E-shaped first part 44 and aplate-shaped second part 43, which form the coil cores. The outer limbsof the first part 44 with an E-shaped cross section are all equal inlength, with the result that they can be closed, for example by adhesivebonding, by the plate-shaped (I-shaped cross section) second part 43without an air gap. In each case one means 81 for influencing theleakage flux is arranged between the central limb and the outer limb ofthe E-shaped part 44 of the coupling means 31. Said means is part of thefirst part 44 and is likewise rectangular and oriented in the same wayas the outer limbs. However, the means 81 for influencing the leakageflux is slightly shorter than the outer limbs, with the result that anair gap 96 is formed with respect to the second part 43 in thepositioned state. The second phase 12 is arranged between the left-handouter limb of the first part 44 and the means 81 for influencing theleakage flux, and the first phase 11 with a current flow in the oppositedirection is arranged between the means 81 and the central limb. Thefirst phase 11 is located between the other side of the central limb andthe further means 81 for influencing the leakage flux. The second phase12 is arranged with a current flow in the opposite direction between thefurther means 81 and the right-hand outer limb of the first part 44. Theouter limbs of the first part 44 have an area A1 in plan view. Thecentral limb of the first part 44 preferably has an area 2*A1 in planview. The limbs of the means 81 for influencing the leakage flux 81 havean area A2. For reasons of simple manufacture, the coupling means 31 to39 are each constructed from the two parts 43, 44, as described. Theouter limb and the central limb are connected to the second part 43 insuch a way that a magnetic circuit is closed. Thus, only small gaps, forexample of the order of magnitude of approximately 10 μm, are permitted.In order that the means 81 build up the desired leakage flux, in theexemplary embodiment the air gap 96 between the ends of the means andthe second part 43 in the positioned state is selected to be of theorder of magnitude of 1 mm, preferably between 0.3 and 0.5 mm. The airgap 96 is also dependent on the geometry of the means 81 for influencingthe leakage flux, in particular on the area A2. Given an area A1 ofapproximately 100 mm2 and an area A2 of likewise 100 mm2, theabove-cited region of the air gap 96 has proven successful.

FIG. 9 now shows a schematic illustration of the matrix-shaped physicaldesign of the concept shown in FIG. 6. As already described inconjunction with FIG. 1, each phase 11 to 16 is magnetically coupled tothree further phases. For this purpose, by way of example for the firstphase 11, three separate coupling means 31, 36, 37 are provided. Thefirst coupling means 31 couples the first phase 11 magnetically to thesecond phase 12. The sixth coupling means 36 couples the first phase 11to the sixth phase 16. The seventh coupling means 37 couples the firstphase 11 to the fourth phase 14. The preferred coupling principle andcoupling means 31 has already been described in conjunction with FIG. 4.The nine separate coupling means 31 to 39 are preferably in the form ofplanar coil cores, for example ferrite cores, which each have twocavities. In each case two conductors or phase sections, physicallyseparated from means 81 to 89 for influencing the leakage flux, of twophases to be coupled are surrounded in these cavities in the couplingmeans 31 to 39, which phases have different current directions in thesesections. The cavities accommodate the two phases (11, 12) to becoupled. However, they could also be at least partially filled with anon-ferromagnetic material.

The plan view in FIG. 9 shows that the phases 11 to 16 have only twoshapes. The first, third and fifth phase 11, 13, 15 is U-shaped. Thesecond, fourth and sixth phase 12, 14, 16 is meandering. In theexemplary embodiment, the phases could be embodied as insulated flexibleround conductors, which are arranged within the coupling means 31 to 39in the same plane.

By way of example, the means 80 to 89 for influencing the leakage fluxhave different shapes. The embodiment shown on the left in FIG. 10 hastwo dome-shaped structures, which are arranged between the phases 11,12. The dome-shaped structures 80 protrude from both sides between thephases 11 and 12, but without touching one another. The ends of themeans 80 for influencing the leakage flux are arranged spaced apart fromone another, with the result that the magnetic circuit is not closed,but the leakage flux is deflected toward the ends. The dome-shapedstructures are part of the second part 44.

The right-hand structure of the means 80 for influencing the leakageflux shown in FIG. 10 has a rectangular cross section, is arrangedbetween the two phases 11, 12 and is part of the second part 44. The endof the means 80 for influencing the leakage flux is oriented in thedirection toward the central limb of the second part 44, but withouttouching it. As a result, no useful magnetic flux is produced. Instead,the leakage flux is guided in a targeted manner by the means 80 and canbe influenced in a targeted manner via the distance or air gap withrespect to the central limb of the second part 44.

The first phase 11 and the second phase 12 are now magnetically coupledto one another by the first coupling means 31. The antiparallel currentconduction indicated results in the resultant magnetic field being keptas low as possible, with the result that the size of the coupling means31 can be minimized. In addition, insulation 45 is provided in each casebetween the first phase 11 and the second phase 12 to electricallyisolate the two phases 11, 12 from one another and in each case withrespect to the coupling means 31.

The exemplary embodiment shown in FIG. 11 differs from that shown inFIG. 10 in that the central limb of the E-shaped first part 44 has anair gap 64 in the direction toward the second part 43. The means 80 forinfluencing the leakage flux are plate-shaped and are arranged betweenthe phases 11, 12. In the exemplary embodiment, the ends of the means 80for influencing the leakage flux are in each case oriented in thedirection of the central limb of the second part 44, without touchingit.

The exemplary embodiments shown in FIGS. 12 to 14 differ from thepreceding exemplary embodiments, in particular in terms of thearrangement of the phases 11, 12. The phases 11, 12 to be coupled are ineach case arranged next to one another and are strip-shaped. In FIG. 12,the means 80 for influencing the leakage flux are dome-shaped.Furthermore, the two sections of the first phase 11 are isolated by thecentral limb of the second part 44, i.e. are arranged adjacent to oneanother.

The exemplary embodiment shown in FIG. 9 differs from that shown in FIG.8 only in terms of the plate-shaped configuration of the means 80 forinfluencing the leakage flux.

The exemplary embodiment shown in FIG. 9 differs from that shown in FIG.8 in that the first and second phases 11, 12 are now always arrangedalternately.

The exemplary embodiment shown in FIG. 11 differs from that shown inFIG. 9 in that the phases 11, 12 are in the form of round conductors.

DESCRIPTION OF THE EXEMPLARY EMBODIMENTS

The described exemplary embodiments function in the manner described inmore detail below. Multiphase converters 10 or DC-to-DC converters withhigh powers without any particular requirements in respect of insulationcan preferably be realized in polyphase arrangements. As a result, thehigh input current I_(E), for example of the order of 300 A, isdistributed among the various six phases 11 to 16 with of the order of50 A in each case. By subsequently superimposing the individual currentsto give an output current IA, a lower AC component can be achieved.Then, the corresponding input or output filters as shown in FIG. 5,illustrated by way of example as capacitors, can be correspondinglysmall. The driving of the phases 11 to 16 takes place sequentially, i.e.successively, with the result that the switch-on times are phase-shiftedin each case through 60 degrees (or temporally through T/6) (in the caseof the six-phase system described), as has been shown already in moredetail in FIG. 11. Depending on the desired voltage ratios, therespective phases 11 to 16 are energized for different durations. Thecorresponding high-side switch of the switching means 21 to 26 is closedfor this purpose. The phases 11 to 16 are not energized when thecorresponding low-side switch of the switching means 21 to 26 is closed.Alternatively, those phases 11 to 16 whose switch-off times are directlypreceding or succeeding could also be considered to be adjacent. Then,the corresponding switch-on points would be selected variably dependingon the desired PWM signal.

In each case one phase 11 is now magnetically coupled to at least threefurther phases 12, 14, 16, to be precise in such a way that the DCcomponents of the individual phases are each compensated for as much aspossible by other phases. This reduces the resultant magnetic field,with the result that the design of the coupling means 31 to 39 or of themagnetic circuit now only needs to be substantially for the magneticfield generated by the AC component. As a result, the coupling means 31to 39, such as coil cores, for example, can be dimensioned so as to becorrespondingly small, which results in considerable savings in terms ofcoupling material, mass and costs. In particular the installation spacecan thus be greatly reduced.

In addition to the two phases which are adjacent in respect of driving(switch-on/switch-off times), the third phase to be coupled is nowpreferably selected in such a way that disruptive mutual influencing ofthe phases is minimized. The selection is performed in such a way thatoptimum compensation of the DC component is achieved. In this case, ithas emerged that, in addition to the adjacent phases (+/−60 degreesphase shift of the switch-on times in the case of six phases; theadjacent phases for the first phase 11 would therefore be the secondphase 12 and the sixth phase 16), the phase with a phase shift of 180degrees (for the first phase 11 this would be the fourth phase 14) isalso particularly suitable since a very high degree of elimination ofthe DC component results there. The two currents through the coupledphases 11, 14 flow in opposition in the seventh coupling means 37. Theresultant current Ires for the magnetization of the coupling means 37 isin this case only triggered by the difference in the currents Ires. TheDC fields cancel one another out to a large extent. The reduced DCcomponent makes itself positively noticeable for the geometry of thecoupling means 31 to 39, which can now manage with a lower volume. Inthe case of six phases 11 to 16, the coupling shown in FIG. 5 has provento be particularly suitable.

Magnetic Coupling

In principle, two phases can be coupled magnetically by virtue of thetwo phases being guided with antiparallel current conduction through arectangular or annular coupling means 31 to 39. It is essential that thecoupling means 31 to 39 is capable of forming a magnetic circuit.

This is possible in the case of a substantially closed structure, whichcan also include an air gap. Furthermore, the coupling means 31 to 39consists of a material conducting a magnetic field with a suitablepermeability.

The coupling concept on which FIG. 6 is based can be explained by way ofexample with reference to FIG. 9. It is essential that the phases to becoupled (in FIG. 8 these are the first phase 11 and the second phase 12)are driven with opposite current flow. The respectively correspondingmagnetic fields cancel one another out substantially in respect of theirDC component, with the result that predominantly only the AC componentnow contributes to the magnetic field generation. As a result, thecorresponding coupling means 31 to 39 can be smaller and it is possibleto dispense with an air gap.

Coupling Means Design

The coupling means 31 to 39 are means for inductive coupling, such as,for example, an iron or ferrite core of a transformer on which thephases 11 to 16 to be coupled generate a magnetic field. The couplingmeans 31 to 39 closes the magnetic circuit of the respective two coupledphases 11 to 16.

The selection of the material for the coupling means 31 to 39 and thepermeability does not play such a significant role for the coupling. Ifno air gap is used, the permeability of the magnetic circuit increases,as a result of which the inductance of the coil becomes greater. As aresult, the current increase becomes flatter and the current waveformscome closer to the ideal direct current. The closer the waveforms cometo a direct current, the lower the resultant current difference betweenthe two phases which are guided (in opposition) through a core ascoupling means 31 to 39. The complexity involved for filters is thusreduced. On the other hand, a system without an air gap has a verysensitive response to different currents between the phases 11 to 16.Although the system is inclined to enter saturation in the case ofrelatively small current faults, it is still quite stable as a result ofthe multiple coupling. In principle, air gaps with different dimensionscan be selected in order to distribute the losses uniformly among thecoupling means 31 to 39. Coupling means 31 to 39 with a lower inductanceL also have, in principle, lower power losses.

In order to arrive at a good compromise between high permeability (smallair gap->less current ripple) and a high degree of robustness (with airgap->high current ripple), different air gaps can be provided. In thisway, the power losses of the coupling means 31 to 39 can also beinfluenced in such a way that desired criteria, for example uniformdistribution of the power losses, are met.

In the exemplary embodiment shown in FIG. 9, the coupling means areprovided with an air gap in one of the diagonals (either coupling means31, 38, 34 or 37, 38, 39). This results in a high level of protectionagainst saturation and, associated therewith, protection againstuncontrolled current increase with only three coupling means 31, 38, 34or 37, 38, 39 with an air gap (which results in a higher current ripple)on all phases 11 to 16. In the case of a large degree of imbalancebetween the phases 11 to 16 or else in the event of failure of aplurality of phases 11 to 16, only individual coupling means 31 to 42would enter saturation, but not all coupling means 31 to 39 of one phaseat a given current.

A further variant would be to design the coupling means 31 to 39 withdifferent air gaps within the structure. The coupling means (in theexemplary embodiment shown in FIGS. 1, 3 and 5, these are the couplingmeans with the reference symbols 37, 38, 39) which are subjected togreater, increased magnetization owing to the driving which isphase-shifted through 180 degrees (as arises as a result of coupling ofthe first phase 11 to the fourth phase 14 by the seventh coupling means37; coupling of the second phase 12 to the fifth phase 15 by the ninthcoupling means 39; coupling of the third phase 13 to the sixth phase 16by the eighth coupling means 38), could be reduced in terms of theirloading, for example, by adaptation or provision of an air gap. Thiswould reduce the total core losses.

In addition, it would be possible in the matrix concept in eachrow/column to provide a coupling means 31 to 39 with a relatively largeair gap or gap. As a result, this coupling means 31 to 39 provided withan air gap would enter saturation first at relatively high currents,with the result that further improved stability in the event of a faultis provided. For reasons of stability, it would be advantageous to guideeach phase 11 to 16 through at least one coupling means 31 to 39, whichenters saturation later than the other coupling means 31 to 39 in thisphase as a result of the provision of a lower inductance L, which couldbe achieved by the provision of an air gap.

The exemplary embodiment shown in FIG. 11 shows an example of a couplingmeans 31 provided with an air gap 64. For this, the central limb of theE-shaped first part 44 is designed to be slightly shorter than the outerlimbs, with the result that an air gap 64 is produced in the directiontoward the second part 43. Alternatively, provision could be made forthe limbs of the E-shaped first part 44 to be equal in size, but for anair gap to be provided between the ends of the limbs and the second part43, for example by means of a nonmagnetic film. Measures which make itpossible to achieve the desired inductance L of the respective couplingmeans 31 to 39, for example by provision of suitable air gap(s) at thesuitable points, are customary to a person skilled in the art.

Design of the Phases

The use of only two geometric shapes of the phases 11 to 16 asillustrated in plan view in FIG. 5 is particularly advantageous in termsof manufacturing technology. One basic shape in this case has a U-shapedprofile. The second basic shape is substantially rectangular ormeandering. The sections shown can be in the form of strip conductors inthe form of leadframes, integrated in corresponding conductor tracks ina printed circuit board or embodied as round conductors.

Further magnetic coupling of the individual cores of the coupling means31 to 39 to form a large total core can result in further savings byvirtue of, for example, a single cover plate 43 being provided for alllower parts 44 of the nine coupling means 31 to 39.

Means 80 to 89 for influencing the leakage flux

In FIG. 12, the useful magnetic flux which passes through theferromagnetic coupling means 43, 44 in each case about two phases 11, 12to be coupled magnetically is indicated by continuous arrows 92. Themagnetic leakage flux is indicated by dashed arrows 94. The magneticleakage flux passes through non-ferromagnetic material, for examplethrough air or a plastic or insulator surrounding the phases 11, 12. Themeans 80 for influencing the leakage flux are now designed in such a waythat they deflect the leakage flux in a targeted manner between twophases 11, 12 to be coupled. This takes place by projections consistingof ferromagnetic material, which are arranged spatially between thephases 11, 12. These projections are connected to the actualferromagnetic coupling means 43, 44 in such a manner as to conductmagnetic flux. The projections can be parts of the coupling means 43,44. However, separate ferromagnetic parts could also be provided.

The means for influencing the leakage flux could have a rectangularcross section. Owing to the particularly simple geometry, such anarrangement can be produced easily and inexpensively.

Alternatively, the means 80 to 89 for influencing the leakage flux canalso have a dome-shaped structure. This is understood to mean, ratherthan a rectangular structure, a structure which tapers toward the end.There could be a continuous, for example parabolic, rounded or circular,transition to the coupling means 44. The domes can be implementeddirectly during the production process (pressing) of the ferrite cores.

The means 80 for influencing the leakage flux is arranged between thephases 11, 12 to be coupled. Said means could protrude only from oneside of the coupling means 44 to the opposite side of the coupling means43, as shown in FIGS. 9 to 11. Alternatively, the means 80 couldprotrude from two sides of the coupling means 43, 44 between the phases11, 12. Preferably, the ends of said means are opposite one another, asindicated in FIG. 8. The ends of the means for influencing the leakageflux are arranged spaced apart from one another, with the result thatthe magnetic circuit is not closed, but the leakage flux is deflectedtoward the ends.

Preferably, the means 80 for influencing the leakage flux are arrangedon the axis of symmetry of two conductors to be coupled. Preferably, thecross section of the means 80 for influencing the leakage field is alsoembodied to be axially symmetrical, in relation to this axis ofsymmetry.

The magnetic leakage flux passes between the ends of the means 80 (FIG.12) or the end of the means and the coupling means 43, 44 (FIGS. 13 to15) in a gap length 86 in a nonferromagnetic medium. The signalcharacteristics of the multiphase converter can be further optimized byvirtue of the geometry of the means 80 and the resultant gap length 86.

As shown in FIG. 15, the phases 11, 12 have now been embodied as roundwire laid next to one another, instead of a flat wire (one above theother: FIG. 10). This increases the leakage fluxes and interference canbe further reduced.

Coupling means 43, 44 and means 80 for influencing a leakage flux havebeen made from the material 3C95. Furthermore, a gap 96 of the order ofmagnitude of 1 mm, for example, has been provided. With this selection,the current ripple/current rate of rise could be further reduced.Saturation effects can be eliminated by virtue of this gap 96.

The described multiphase converter 10 is particularly suitable for usein a motor vehicle electrical distribution system, in which inparticular dynamic load requirements are of subordinate importance. Inparticular for such comparatively sluggish systems, the described designis suitable.

In the core model used at present, the leakage flux is set by anadditional leakage limb, which is introduced between the two turns.Owing to the design of the core, the response can be adjustedindividually to the application. Main parameters are in this case thetwo air gaps which can be defined in an application-specific manner.

By virtue of the introduction of more than one turn per coupled phase,the core losses which arise as a result of the remagnetization of thecore can be greatly reduced. In this case, turns numbers of 2 or 3, orat least in the low range, are expedient in order to keep the turnslosses owing to the winding length low.

In the case of the coupling means shown in the figures, more than twoturns per phase can be provided.

1. A multiphase converter comprising a plurality of electrical phases(11 to 16), which are each drivable by one of a plurality of switches(21 to 26), wherein at least one coupler (31 to 39; 100 to 106; 150 to156) for coupling a phase (11 to 16) to a further phase (11 to 16) isprovided, wherein two turns (110 to 116; 120; 122; 130; 132) areprovided per phase (11 to 16).
 2. The multiphase converter as claimed inclaim 1, wherein the multiphase converter is configured for at least twophases.
 3. The multiphase converter as claimed in claim 1, in which atleast one of the at least one coupler (31 to 39; 100 to 106; 150 to 156)has at least one means (80 to 89) for influencing a magnetic leakageflux.
 4. The multiphase converter as claimed in claim 2, in which themeans (80 to 89) for influencing the leakage flux is connected either ononly one side or on two sides to the at least one coupler (31 to 39; 100to 106; 150 to 156).
 5. The multiphase converter as claimed in claim 2,characterized in that a gap is provided between the means (80 to 89) forinfluencing the leakage flux and the at least one coupler (31 to 39; 100to 106; 150 to 156).
 6. The multiphase converter as claimed in claim 2,in which the means (80 to 89) for influencing the leakage flux isrectangular.
 7. The multiphase converter as claimed in claim 1, in whichat least one phase (11 to 16) is U-shaped and the phase (11 to 16)coupled thereto is meandering.
 8. The multiphase converter as claimed inclaim 1, in which the plurality of switches (21 to 26) drive the phases(11 to 16) sequentially, and the first phase (11 to 16) is magneticallycoupled to at least one further phase (11 to 16), which is drivendirectly beforehand and afterward.
 9. The multiphase converter asclaimed in claim 1, in which a phase (11 to 16) is magnetically coupledto at least one further phase (11 to 16), which is driven with a phaseshift substantially through approximately 180°.
 10. The multiphaseconverter as claimed in claim 2, in which the means (80 to 89) forinfluencing the leakage flux is dome-shaped.